The present invention relates generally to a circuit for driving a reactive load, and more particularly, to a highly efficient resonant switching circuit for converting DC current into sinusoidal circulating currents in reactive loads at radio frequencies. The present invention can be used, for instance, for driving reactive (inductive) loop antennas such as that used in an interrogator for an electronic article surveillance (EAS) system.
A drive circuit with a resonant circuit is commonly used to enable the efficient conversion of energy from a DC power supply to a reactive load. FIG. 1 shows, in generalized form, a prior art drive circuit 100 for driving a reactive (inductive) load 102 (Ls). The drive circuit 100 includes a current switch device Qs, a resonance capacitor (Cs) and loss element (Ro), the latter representing the power losses associated with the resistances of the reactive load Ls 102 and the capacitor Cs and any additional resistance that may be connected to the circuit 100. The design of the circuit 100 is optimized for delivering power into the loss element (Ro), rather than reactive energy into the inductive load (Ls). Thus, the analysis of the efficiency of the circuit 100 is commonly relative to the amount of power delivered to the loss element (Ro). The following discussion refers to this common method of analysis. (An additional resistance may be made a part of the resonant circuit comprising Ls and Cs, for example, to increase the resonance bandwidth).
FIG. 2 shows voltage and current waveforms 102, 104 typically associated with the drive circuit 100. The upper waveform 104 shows the voltage (Vs) across the current switch device Qs and the capacitor Cs resulting from the current switching performed by the current switch device Qs. The lower waveform 106 shows the current (Ils) that flows through the reactive load Ls.
It is desirable to operate drive circuits for reactive loads with the highest possible efficiency. Inefficient drive circuits require larger power supplies. Inefficient drive circuits also waste substantial power in the form of heat, and thus require large heat sinks and/or cooling fans for heat removal, and are often less reliable. The nature of the current switch device Qs determines the efficiency of the prior art drive circuit 100. In particular, the percentage of the time the switch device Qs is made to operate in the linear mode, a mode where the current is made to vary as a continuous function of time instead of an on/off function of time, determines the so called class of operation of the prior art drive circuit 100.
In reactive load driver circuits, such as the drive circuit 100, the power conversion efficiency is generally referred to as the amount of power dissipated by the loss element Ro (the resistive losses of the circuit). Thus, the power conversion efficiency is the percentage of the power dissipated in Ro divided by the total power consumed by the drive circuit 100 (the sum of the power delivered to Ro and the power dissipated by current switch device Qs).
Commonly known classes of operation of the drive circuit 100 are Class A, Class B and Class C. Class A operation refers to operating Qs in the linear mode 100% of the time. Class A operation is very inefficient because of the power dissipated across the current switch device Qs. This power dissipation is caused by the simultaneous voltage across and current flow through the current switch device Qs, that results from the linear mode of operation of Qs. Class A operation of the prior art drive circuit 100 has a theoretical maximum efficiency of 25%.
Class B operation of the circuit 100 refers to operating the current switch device Qs in the linear mode for about 50% of the time. In other words, the switch device Qs is made to operate linearly for about one half of each cycle of the drive waveform. The maximum theoretical power conversion efficiency for Class B operation of the prior art circuit 100 is 78.65%, although practical implementations often achieve less than 50% efficiency.
Class C operation of the circuit 100 refers to operating the current switch device Qs in the linear mode for less than 50% of the time. In fact, Class C operation of the circuit 100 may operate the current switch device Qs predominantly as an on/off switch, thus not making it suitable for true linear amplification applications. The conduction time diagram shown in FIG. 2 is for Class C operation. Class C operation of the prior art circuit 100 achieves the highest efficiency operation, often between 40% and 80% in practical applications. Such efficiencies still do not fulfill the objective of the present invention.
FIG. 3 shows a prior art "flyback" drive circuit 108, commonly used as a horizontal deflection drive circuit in CRT displays (televisions and monitors). When used as a deflection drive circuit in CRT's, the drive circuit 108 includes a high voltage transformer (Ls), a current switching device (Qs), and a resonance capacitor (Cs). The drive circuit 108 may also include a large value coupling capacitor (Cc), to prevent DC current from flowing through the deflection coil (Lo) inductance that would cause horizontal positioning errors in the CRT display.
The drive circuit 108 may be characterized as a resonant switching drive circuit because the current switching device Qs is operated strictly in the on/off mode. The resonant part of the drive circuit 108 is formed by the parallel combination of the deflection coil (Lo) and the high voltage transformer (Ls) in conjunction with the resonance capacitor (Cs). When operated as a horizontal deflection circuit, the current switching device Qs is closed for the sweep duration (about 80% of the total period), causing a flat bottomed voltage waveform to be applied across the deflection coil (Lo). (See waveforms Vs and Vo in FIG. 3). During the time that the current switching device Qs is on, the supply voltage (Vsp) is applied across the inductors (Ls) and (Lo). As is well known in the art, the currents that flow through Ls and Lo increase linearly during this time. This linear current increase is desirable in that it causes a more or less linear deflection of the electrons of the CRT as a function of time, thereby causing a more or less uniform distribution of information across the screen of the CRT.
When the switching device Qs opens during the so called flyback time (about 20% of the total period), the energy stored in the inductors Ls and Lo is transferred in resonant fashion to the resonance capacitor (Cs). This results in the generation of the high voltage half sinusoid signal across the capacitor (Cs), the peak of which is much higher in amplitude than the power supply voltage (Vsp). Thus, the voltage across the inductors Ls and Lo is reversed, as compared to the voltage applied across them when the current switching device Qs was closed, thereby causing the current flowing through them to reverse, which in turn, causes the capacitor (Cs) to discharge and transfer its stored energy back to the combination of inductors Ls and Lo. This charge and discharge of the capacitor (Cs) is known as flyback and occurs in a sinusoidal manner, thus resulting in the half-sine flyback pulses that are indicative of the operation of the drive circuit 108.
The flyback drive circuit 108 converts DC power to reactive energy at RF frequencies very efficiently. Since the current switching device (Qs) is used as a switch, and not as a linear device, the power losses associated with Qs can be very low. Unfortunately, the flyback drive circuit 108 is not suitable for driving an inductive loop antenna because of the high harmonic content of the signal it generates. These harmonics radiate, thereby creating a high level of emissions outside of the frequency range of the intended radiation, which is unacceptable to government radio regulation authorities, such as the U.S. Federal Communications Commission.
FIG. 4 shows a prior art Class E drive circuit 110 for driving an inductive load (Lo). The circuit 110 includes a current switching device (Qs), a switch capacitor (Cs), a DC feed inductor (Ls), a resonance capacitor (Co), the output inductor (Lo) (which may be an inductive loop antenna), and a loss element (Ro), the latter representing the power losses associated with the resistances of Ls, Cs, Co, Lo and any additional resistance that may be connected to the circuit 110. (As with the circuit 100 of FIG. 1, an additional resistance may be made a part of the resonant circuit comprising Lo and Co, for example, to increase the resonance bandwidth).
FIG. 5 shows the voltage and current waveforms associated with the Class E drive circuit 110. A half-sine flyback pulse 112 is produced at the switching device Qs by the switch capacitor (Cs), the output inductor (Lo) and the resonance capacitor (Co). A distinguishing feature of Class E drive circuit 110 is that the AC component of the current (Ils) 114 in the switch inductor (Ls) is much smaller than the DC current 116 flowing through the switch inductor (Ls).
In the Class E drive circuit 110, the current switching device Qs is operated as a switch, either on or off. When on, the current switching device Qs conducts for the low voltage portion of the half sine wave and therefore, minimum power is dissipated. When off, no current flows through the current switching device Qs, and therefore essentially no power is dissipated. In the Class E drive circuit 110, the DC feed inductor Ls has a large value relative to the output inductor Lo, and therefore does not affect the resonance operation of the circuit 110. The resonant frequency of the output inductor Lo and the resonance capacitor Co is chosen to be nominally at Fo, the switching frequency of the current switching device Qs. This is so that the resonant circuit comprising Lo and Co filters out the harmonics of the half sine signal generated across the switch Qs, thereby ensuring that the radiated signal output from the inductor Lo is mostly free of unwanted harmonics. The half sine portion of the signal Vs shown in FIG. 5 is the result of the combined action of Cs, Co and Lo.
In a practical implementation of the Class E driver circuit 110, the resonant frequency of Cs, Co and Lo may be slightly higher than the operating frequency Fo. This is to ensure that signal Vs returns to ground before the current switch Qs is turned on. This minimizes the power losses from the current switch Qs associated with switching. We have determined that a practical implementation of the Class E driver circuit for use as a loop antenna driver is unsuitable because a practical switching device Qs comprises an FET that has a large, non-linear device capacitance. This device capacitance is at maximum when the voltage across the device (Vs) is minimum. In practice, this large non-linear device capacitance causes the resonance frequency of the circuit to be dramatically lower during the immediate period after the FET is turned off. This tends to latch the circuit such that the drive voltage (Vs) is held low long after the FET is turned off. This latching effect can last for more than one cycle, until the current that flows through the DC feed inductor (Ls) increases sufficiently to charge the large non-linear capacitance of the FET sufficiently to pull the circuit out of this state. Thus, in a practical implementation of the Class E driver circuit 110, drive signal cycles may be skipped, due to latching, either periodically (generating a sub-harmonic signal) or randomly (generating a chaotic form of noise). Thus, a practical implementation of the Class E driver circuit 110 is not suitable for use as a driver for a reactive load such as a loop antenna.
Class A, B and C and flyback drivers are more immune to such problems because the resonance of these circuits controls their operation to a much greater extent than that of the Class E circuit. The inductor Ls in the Class A, B and C drive circuits 100 of FIG. 1 and the flyback drive circuit 108 of FIG. 3 is relatively much smaller in value than the inductor Ls of the Class E drive circuit 110. With a relatively small value of Ls, the current increase through Ls (associated with the applied voltage across it when the current switch Qs is conducting) charges the non-linear capacitance of practical switching devices Qs (such as an FET) sufficiently quickly so that the previously described latching does not occur.
However, circuits using these classes (A, B, C) of operation are either inefficient or generate unacceptable harmonics. Despite the availability of many different types of driver circuits, there is still a need for a driver circuit that can efficiently drive reactive loads without the introduction of excessive noise or harmonics and which is suitable for driving an inductive loop antenna. The present invention fulfills such needs.